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The LM105-An Improved Positive Regulator National Semiconductor Application Note 23 January 1969 Robert J. Widlar Apartado Postal 541 Puerto Vallarta, Jalisco Mexico introduction IC voltage regulators are seeing rapidly increasing usage. The LM100, one of the first, has already been widely ac- cepted. Designed for versatility, this circuit can be used as a linear regulator, a switching regulator, a shunt regulator, or even a current regulator. The output voltage can be set be- tween 2V and 30V with a pair of external resistors, and it works with unregulated input voltages down to 7V. Dissipa- tion limitations of the IC package restrict the output current to less than 20 mA, but external transistors can be added to obtain output currents in excess of 5A. The LM100 and an extensive description of its use in many practical circuits are described in References 1±3. One complaint about the LM100 has been that it does not have good enough regulation for certain applications. In ad- dition, it becomes difficult to prove that the load regulation is satisfactory under worst-case design conditions. These problems prompted development of the LM105, which is nearly identical to the LM100 except that a gain stage has been added for improved regulation. In the great majority of applications, the LM105 is a plug-in replacement for the LM100. the improved regulator The load regulation of the LM100 is about 0.1%, no load to full load, without current limiting. When short circuit protec- tion is added, the regulation begins to degrade as the output current becomes greater than about half the limiting current. This is illustrated inFigure1. The LM105, on the other hand, gives 0.1% regulation up to currents closely approaching the short circuit current. As shown inFigure1b, this is partic- ularly significant at high temperatures. The current limiting characteristics of a regulator are impor- tant for two reasons: First, it is almost mandatory that a regulator be short-circuit protected because the output is distributed to enough places that the probability of it becom- ing shorted is quite high, Secondly, the sharpness of the limiting characteristics is not improved by the addition of external booster transistors. External transistors can in- crease the maximum output current, but they do not im- prove the load regulation at currents approaching the short TL/H/6906±1 a. T j e 25 § C TL/H/6906±2 b. T j e 125 § C Figure 1. Comparison between the load regulation of the LM100 and LM105 for equal short circuit currents circuit current. Thus, it can be seen that the LM105 provides more than ten times better load regulation in practical power supply designs. C 1995 National Semiconductor Corporation TL/H/6906 RRD-B30M115/Printed in U. S. A. Figure 2 shows that the LM105 also provides better line regulation than the LM100. These curves give the percent- age change in output voltage for an incremental change in the unregulated input voltage. They show that the line regu- lation is worst for small differences between the input and output voltages. The LM105 provides about three times bet- ter regulation under worst case conditions. Bypassing the internal reference of the regulator makes the ripple rejection of the LM105 almost a factor of ten better than the LM100 over the entire operating range, as shown in the figure. This bypass capacitor also eliminates noise generated in the in- ternal reference zener of the IC. TL/H/6906±3 Figure 2. Comparison between the line regulation char- acteristics of the LM100 and LM105 TL/H/6906±5 Figure 4. Schematic diagram of the LM105 regulator Q4, divided down by R1 and R2 and connected in series with a diode-connected transistor, Q7. The negative temper- ature coefficient of Q7 cancels out the positive coefficient of the voltage across R2, producing a temperature-compen- sated 1.8V on the base of Q8. This point is also brought outside the circuit so that an external capacitor can be add- ed to bypass any noise from the zener diode. Transistors Q8 and Q9 make up the error amplifier of the circuit. A gain of 2000 is obtained from this single stage by using a current source, another collector on Q2, as a collec- tor load. The output of the amplifier is buffered by Q11 and used to drive the series-pass transistor, Q12. The collector of Q12 is brought out so that an external PNP transistor, or PNPÐNPN combination, can be added for increased output current. Current limiting is provided by Q10. When the voltage across an external resistor connected between Pins 1 and 8 becomes high enough to turn on Q10, it removes the base drive from Q11 so the regulator exhibits a constant-current characteristic. Prebiasing the current limit transistor with a portion of the emitter-base voltage of Q12 from R6 and R7 reduces the current limit sense voltage. This increases the The LM105 has also benefited from the use of new IC com- ponents developed after the LM100 was designed. These have reduced the internal power consumption so that the LM105 can be specified for input voltages up to 50V and output voltages to 40V. The minimum preload current re- quired by the LM100 is not needed on the LM105. circuit description The differences between the LM100 and the LM105 can be seen by comparing the schematic diagrams inFigures3and 4. Q4 and Q5 have been added to the LM105 to form a common-collector, common-base, common-emitter amplifi- er, rather than the single common-emitter differential ampli- fier of the LM100. In the LM100, generation of the reference voltage starts with zener diode, D1, which is supplied with a fixed current from one of the collectors of Q2. This regulated voltage, which has a positive temperature coefficient, is buffered by TL/H/6906±4 Figure 3. Schematic diagram of the LM100 regulator 2 efficiency of the regulator, especially when foldback current limiting is used. With foldback limiting, the voltage dropped across the current sense resistor is about four times larger than the sense voltage. As for the remaining details, the collector of the amplifier, Q9, is brought out so that external collector-base capaci- tance can be added to frequency-stabilize the circuit when it is used as a linear regulator. This terminal can also be grounded to shut the regulator off. R9 and R4 are used to start up the regulator, while the rest of the circuitry estab- lishes the proper operating levels for the current source transistor, Q2. The reference circuitry of the LM105 is the same, except that the current through the reference divider, R2, R3 and R4, has been reduced by a factor of two on the LM105 for reduced power consumption. In the LM105, Q2 and Q3 form an emitter coupled amplifier, with Q3 being the emitter-fol- lower input and Q2 the common-base output amplifier. R6 is the collector load for this stage, which has a voltage gain of about 20. The second stage is a differential amplifier, using Q4 and Q5. Q5 actually provides the gain. Since it has a current source as a collector load, one of the collectors of Q12, the gain is quite high: about 1500. This gives a total gain in the error amplifier of about 30,000 which is ten times higher than the LM100. It is not obvious from the schematic, but the first stage (Q2 and Q3) and second stage (Q4 and Q5) of the error amplifi- er are closely balanced when the circuit is operating. This will be true regardless of the absolute value of components and over the operating temperature range. The only thing affecting balance is component matching, which is good in a monolithic integrated circuit, so the error amplifier has good drift characteristics over a wide temperature range. Frequency compensation is accomplished with an external integrating capacitor around the error amplifier, as with the LM100. This scheme makes the stability insensitive to load- ing conditionsÐresistive or reactiveÐwhile giving good transient response. However, an internal capacitor, C1, is added to prevent minor-loop oscillations due to the in- creased gain. Additional differences between the LM100 and LM105 are that a field-effect transistor, Q18, connected as a current source starts the regulator when power is first applied. Since this current source is connected to ground, rather than the output, the minimum load current before the regula- tor drops out of operation with large input-output voltage differentials is greatly reduced. This also minimizes power dissipation in the integrated circuit when the difference be- tween the input and output voltage is at the worst-case val- ue. With the LM105 circuit configuration, it was also neces- sary to add Q17 to eliminate a latch-up mechanism which could exist with lower output-voltage settings. Without Q17, this could occur when Q3 saturated and cut off the second stage amplifiers, Q4 and Q5, causing the output to latch at a voltage nearly equal to the unregulated input. power limitations Although it is desirous to put as much of the regulator as possible on the IC chip, there are certain basic limitations. For one, it is not a good idea to put the series pass transis- tor on the chip. The power that must be dissipated in the pass transistor is too much for practical IC packages. Fur- ther, IC's must be rated at a lower maximum operating tem- perature than power transistors. This means that even with a power package, a more massive heat sink would be re- quired if the pass transistor was included in the IC. Assuming that these problems could be solved, it is still not advisable to put the pass transistor on the same chip with the reference and control circuitry: changes in the unregu- lated input voltage or load current produce gross variations in chip temperature. These variations worsen load and line regulation due to temperature interaction with the control and reference circuitry. To elaborate, it is reasonable to neglect the package prob- lem since it is potentially solvable. The lower, maximum op- erating temperatures of IC's, however, present a more basic problem. The control circuitry in an IC regulator runs at fairly low currents. As a result, it is more sensitive to leakage currents and other phenomena which degrades the per- formance of semiconductors at high temperatures. Hence, the maximum operating temperature is limited to 150 § Cin military temperature range applications. On the other hand, a power transistor operating at high currents may be run at temperatures up to 200 § C, because evena1mAleakage current would not affect its operation in a properly designed circuit. Even if the pass transistor developed a permanent 1 mA leakage from channeling, operating under these con- ditions of high stress, it would not affect circuit operation. These conditions would not trouble the pass transistor, but they would most certainly cause complete failure of the con- trol circuitry. These problems are not eliminated in applications with a lower maximum operating temperature. Integrated circuits are sold for limited temperature range applications at con- siderably lower cost. This is mainly based on a lower maxi- mum junction temperature. They may be rated so that they do not blow up at higher temperatures, but they are not guaranteed to operate within specifications at these temper- atures. Therefore, in applications with a lower maximum am- bient temperature, it is necessary to purchase an expensive full temperature range part in order to take advantage of the theoretical maximum operating temperatures of the IC. Figure5makes the point about dissipation limitations more strongly. It gives the maximum short circuit output current for an IC regulator in a TO-5 package, assuming a 25 § C temperature rise between the chip and ambient and a quies- cent current of 2 mA. Dual-in-line or flat packages give re- sults which are, at best, slightly better, but are usually worse. If the short circuit current is not of prime concern, Figure5 can also be used to give the maximum output cur- rent as a function of input-output voltage differential, How- ever, the increased dissipation due to the quiescent current flowing at the maximum input voltage must be taken into account. In addition, the input-output differential must be measured with the maximum expected input voltages. 3 TL/H/6906±6 Figure 5. Dissipation limited short circuit output current for an IC regulator in a TO-5 package The 25 § C temperature rise assumed in arriving at Figure5 is not at all unreasonable. With military temperature range parts, this is valid for a maximum junction temperature of 150 § C with a 125 § C ambient. For low cost parts, marketed for limited temperature range applications, this maximum differential appropriately derates the maximum junction tem- perature. In practical designs, the maximum permissible dissipation will always be to the left of the curve shown for an infinite heat sink inFigure5. This curve is realized with the package immersed in circulating acetone, freon or mineral oil. Most heat sinks are not quite as good. To summarize, power transistors can be run with a tempera- ture differential, junction to ambient, 3 to 5 times as great as an integrated circuit. This means that they can dissipate much more power, even with a smaller heat sink. This, cou- pled with the fact that low cost, multilead power packages are not available and that there can be thermal interactions between the control circuitry and the pass transistor, strong- ly suggests that the pass transistors be kept separate from the integrated circuit. using booster transistors Figure6 shows how an external pass transistor is added to the LM105. The addition of an external PNP transistor does not increase the minimum input output voltage differential. This would happen if an NPN transistor was used in a com- pound emitter follower connection with the NPN output tran- sistor of the IC. A single-diffused, wide base transistor like the 2N3740 is recommended because it causes fewer oscil- lation problems than double-diffused, planar devices. In ad- dition, it seems to be less prone to failure under overload conditions; and low cost devices are available in power packages like the TO-66 or even TO-3. When the maximum dissipation in the pass transistor is less than about 0.5W, a 2N2905 may be used as a pass transis- tor. However, it is generally necessary to carefully observe thermal deratings and provide some sort of heat sink. In the circuit of Figure6, the output voltage is determined by R1 and R2. The resistor values are selected based on a feedback voltage of 1.8V to Pin 6 of the LM105. To keep thermal drift of the output voltage within specifications, the parallel combination of R1 and R2 should be approximately 2K. However, this resistance is not critical. Variations of g 30% will not cause an appreciable degradation of temper- ature drift. The 1 m F output capacitor, C2, is required to suppress oscil- lations in the feedback loop involving the external booster transistor, Q1, and the output transistor of the LM105. C1 compensates the internal regulator circuitry to make the sta- bility independent for all loading conditions. C3 is not nor- mally required if the lead length between the regulator and the output filter of the rectifier is short. Current limiting is provided by R3. The current limit resistor should be selected so that the maximum voltage drop across it, at full load current, is equal to the voltage given in Figure 7 at the maximum junction temperature of the IC. This assures a no load to full load regulation better than 0.1% under worst-case conditions. TL/H/6906±7 TL/H/6906±8 Figure 7. Maximum voltage drop across current limit re- sistor at full load for worst case load regula- tion of 0.1% The short circuit output current is also determined by R3. Figure8 shows the voltage drop across this resistor, when the output is shorted, as a function of junction temperature in the IC. With the type of current limiting used inFigure6, the dissipa- tion under short circuit conditions can be more than three times the worst-case full load dissipation. Hence, the heat Figure 6. 0.2A regulator 4 the current is reduced to a value determined by the current limit resistor and the current limit sense voltage of the LM105. TL/H/6906±9 Figure 8. Voltage drop across current limit resistor re- quired to initiate current limiting sink for the pass transistor must be designed to accommo- date the increased dissipation if the regulator is to survive more than momentarily with a shorted output. It is encourag- ing to note, however, that the short circuit current will de- crease at higher ambient temperatures. This assists in pro- tecting the pass transistor from excessive heating. foldback current limiting With high current regulators, the heat sink for the pass tran- sistor must be made quite large in order to handle the power dissipated under worst-case conditions. Making it more than three times larger to withstand short circuits is sometimes inconvenient in the extreme. This problem can be solved with foldback current limiting, which makes the output cur- rent under overload conditions decrease below the full load current as the output voltage is pulled down. The short cir- cuit current can be made but a fraction of the full load cur- rent. A high current regulator using foldback limiting is shown in Figure9. A second booster transistor, Q1, has been added to provide 2A output current without causing excessive dis- sipation in the LM105. The resistor across its emitter base junction bleeds off any collector base leakage and estab- lishes a minimum collector current for Q2 to make the circuit easier to stabilize with light loads. The foldback characteris- tic is produced with R4 and R5. The voltage across R4 bucks out the voltage dropped across the current sense resistor, R3. Therefore, more voltage must be developed across R3 before current limiting is initiated. After the output voltage begins to fall, the bucking voltage is reduced, as it is proportional to the output voltage. With the output shorted, TL/H/6906±11 Figure 10. Limiting characteristics of regulator using foldback current limiting Figure10 illustrates the limiting characteristics. The circuit regulates for load currents up to 2A. Heavier loads will cause the output voltage to drop, reducing the available cur- rent. With a short on the output, the current is only 0.5A. In design, the value of R3 is determined from R 3 e V lim I SC , (1) where V lim is the current limit sense voltage of the LM105, given in Figure8, and I SC is the design value of short circuit current. R5 is then obtained from R 5 e V OUT a V sense I bleed a I bias , (2) where V OUT is the regulated output voltage, V sense is maxi- mum voltage across the current limit resistor for 0.1% regu- lation as indicated in Figure7,I bleed is the preload current on the regulator output provided by R5 and I bias is the maxi- mum current coming out of Pin 1 of the LM105 under full load conditions. I bias will be equal to 2 mA plus the worst- case base drive for the PNP booster transistor, Q2. I bleed should be made about ten times greater than I bias . Finally, R4 is given by R 4 e I FL R 3 b V sense I bleed , (3) where I FL is the output current of the regulator at full load. ² Solid tantalum ³ Ferroxcube K5-001-00/3B TL/H/6906±10 Figure 9. 2A regulator with foldback current limiting 5 [ Pobierz całość w formacie PDF ] |
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